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SWITCHMODEt Power
Reference Manual and Des
Rev
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SMPSRM
ON Semiconductor and are registered trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC re
changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitabparticular purpose, nor does SCILLC assume any liability arising out of the application or use of any product or circuit, and specificliability, including without limitation special, consequential or incidental damages. Typical parameters which may be provided in Sspecifications can and do vary in different applications and actual performance may vary over time. All operating parameters, incvalidated for each customer application by customers technical experts. SCILLC does not convey any license under its patent righSCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the bintended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation whermay occur. Should Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indand its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even
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Forward
Every new electronic product, except those that are battery powered, requires con
115 Vac or 230 Vac power to some dc voltage for powering the electronics. The ava
and application information and highly integrated semiconductor control ICs for
supplies allows the designer to complete this portion of the system design qui
Whether you are an experienced power supply designer, designing your first s
supply or responsible for a make or buy decision for power supplies, the variety
in the SWITCHMODE Power Supplies Reference Manual and Design Guid
useful.
ON Semiconductor has been a key supplier of semiconductor products for switchin
since we introduced bipolar power transistors and rectifiers designed specifical
power supplies in the mid70s. We identified these as SWITCHMODE produ
power supply designed using ON Semiconductor components can rightful
SWITCHMODE power supply or SMPS.
This brochure contains useful background information on switching power suppliwant to have more meaningful discussions and are not necessarily experts on powe
provides real SMPS examples, and identifies several application notes and a
resources available from ON Semiconductor, as well as helpful books availab
publishers and useful web sites for those who are experts and want to increase th
extensive list and brief description of analog ICs, power transistors, rectifiers an
components available from ON Semiconductor for designing a SMPS are also
includes our newest GreenLine, Easy Switcher and very high voltage ICs (VHV
high efficiency HDTMOS and HVTMOS power FETs, and a wide choice of d
in surface mount packages.
For the latest updates and additional information on analog and discrete products for p
power management applications, please visit our website: (http://onsemi.com).
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SMPSRM
Table of Contents
Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Linear versus Switching Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Switching Power Supply Fundamentals . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
The ForwardMode Converter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
The FlybackMode Converter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Common Switching Power Supply Topologies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Interleaved Multiphase Converters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Selecting the Method of Control . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . The Choice of Semiconductors . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Power Switches . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
The Bipolar Power Transistor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
The Power MOSFET . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Driving MOSFETs in Switching Power Supply Applications . . . . . . . . . . . . . . . . . . . . . . . . . . .
The Insulated Gate Bipolar Transistor (IGBT) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Rectifiers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
The Magnetic Components . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Laying Out the Printed Circuit Board . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Losses and Stresses in Switching Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Techniques to Improve Efficiency in Switching Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . .
The Synchronous Rectifier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Snubbers and Clamps . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
The Lossless Snubber . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
The Active Clamp . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
QuasiResonant Topologies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Power Factor Correction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
SMPS Examples . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Integrated Circuits for Switching Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Suggested Components for Specific Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Literature Available from ON Semiconductor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Application Notes, Brochures, Device Data Books and Device Models . . . . . . . . . . . . . . . . . .
References for Switching Power Supply Design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Books . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Websites . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Analog ICs for SWITCHMODE Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
ON S mi d t W ld id S l Offi
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IntroductionThe neverending drive towards smaller and lighter
products poses severe challenges for the power supply
designer. In particular, disposing of excess heat
generated by power semiconductors is becoming more
and more difficult. Consequently it is important that the
power supply be as small and as efficient as possible, and
over the years power supply engineers have responded to
these challenges by steadily reducing the size and
improving the efficiency of their designs.
Switching power supplies offer not only higher
efficiencies but also greater flexibility to the designer.
Recent advances in semiconductor, magnetic and passive
technologies make the switching power supply an ever
more popular choice in the power conversion arena.
This guide is designed to give the prospective designer
an overview of the issues involved in designing
switchmode power supplies. It describes the basic
operation of the more popular topologies of switching
power supplies, their relevant parameters, provides
circuit design tips, and information on how to select the
most appropriate semiconductor and passive
components. The guide also lists the ON Semiconductor
components expressly built for use in switching power
supplies.
Linear versus SwitchingPower Supplies
Switching and linear regulators use fundamentally
different techniques to produce a regulated outputvoltage from an unregulated input. Each technique has
advantages and disadvantages, so the application will
determine the most suitable choice.
Linear power supplies can only stepdown an input
voltage to produce a lower output voltage. This is done
by operating a bipolar transistor or MOSFET pass unit in
its linearoperating mode; that is, the drive to the pass unit
is proportionally changed to maintain the required output
voltage. Operating in this mode means that there isalways a headroom voltage, Vdrop, between the input
and the output. Consequently the regulator dissipates a
considerable amount of power, given by (Vdrop Iload).
This headroom loss causes the linear regulator to only
be 35 to 65 percent efficient. For example, if a 5.0 V
regulator has a 12 V input and is supplying 100 mA, it
A low dropout (LDO) regulato
output stage that can reduce Vdrop
than 1.0 V. This increases the effici
linear regulator to be used in higher
Designing with a linear regulator
requiring few external components
considerably quieter than a switch
highfrequency switching noise.
Switching power supplies operate
the pass units between two efficien
cutoff, where there is a high voltage
but no current flow; and saturation,
current through the pass unit but at a
drop. Essentially, the semicondu
creates an AC voltage from the inp
AC voltage can then be steppe
transformers and then finally filtere
output. Switching power supplie
efficient, ranging from 65 to 95 perc
The downside of a switching d
considerably more complex. In a
voltage contains switching noise
removed for many applications.
Although there are clear differen
and switching regulators, many appli
types to be used. For example, a swit
provide the initial regulation, then a
provide postregulation for a noise
design, such as a sensor interface ci
Switching Power SupplyFundamentals
There are two basic types of puls
(PWM) switching power supplies,
boostmode. They differ in the
elements are operated. Each basic typ
and disadvantages.
The ForwardMode ConverterThe forwardmode converter can bpresence of an LC filter on its out
creates a DC output voltage, whic
volttime average of the LC
rectangular waveform. This can be e
V t[ Vi @ duty cy
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SMPSRM
Ipk
TIME
IloadImin
PowerSwitch
OFF
PowerSwitch
OFFPowerSwitch
ON
PowerSwitch
ON
Vsat
Power SW Power SW DiodeDiodeTIME
Figure 1. A Basic ForwardMode Converter and Waveforms (Buck Converter Sh
Vfwd
INDUCTORCURRENT
(AMP
S)
DIODEVOLTA
GE
(VOLTS)
LO
RloadCout
DVin
SW
Ion Ioff
Its operation can be better understood when it is broken
into two time periods: when the power switch is turned
on and turned off. When the power switch is turned on,the input voltage is directly connected to the input of the
LC filter. Assuming that the converter is in a
steadystate, there is the output voltage on the filters
output. The inductor current begins a linear ramp from an
initial current dictated by the remaining flux in the
inductor. The inductor current is given by:
iL(on)+(Vin* Vout)
Lt) iinit 0v tv ton (eq. 2)
During this period, energy is stored as magnetic flux
within the core of the inductor. When the power switch
is turned off, the core contains enough energy to supply
the load during the following off period plus some
reserve energy.
When the power switch turns off, the voltage on the
clamped when the catch diode D
biased. The stored energy then cont
output through the catch diode aninductor current decreases from an in
given by:
iL(off)+ ipk*Voutt
L0v tv
The off period continues until the
power switch back on and the cycle
The buck converter is capable of o
output power, but is typically used foapplications whose output powers are
Compared to the flybackmode con
converter exhibits lower output p
voltage. The disadvantage is that
topology only. Since it is not an iso
safety reasons the forward converte
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The FlybackMode Converter
The basic flybackmode converter uses the same
components as the basic forwardmode converter, but in
a different configuration. Consequently, it operates in a
different fashion from the forwardm
most elementary flybackmode con
stepup converter, is shown in Figu
Figure 2. A Basic BoostMode Converter and Waveforms (Boost Converter Sh
PowerSwitch
ON
Vin
Power
Switch
ON
Diode
ON
Vflbk(Vout)
Diode
ON
Power
Switch
ON
Ipk
INDUC
TORCURRENT
(AMPS)
SWITCH
VOLTAGE
(VOLTS)
Iload
Vsat
L
Rload
Cout
D
VinIoff
SWIloadIon
Again, its operation is best understood by considering the
on and off periods separately. When the power
switch is turned on, the inductor is connected directly
across the input voltage source. The inductor current then
rises from zero and is given by:
iL(on)+Vint
Lv tv 0on (eq. 4)
Energy is stored within the flux in the core of the inductor.
The peak current, ipk, occurs at the instant the power
switch is turned off and is given by:
the output rectifier when its voltage
voltage. The energy within the core o
passed to the output capacitor. Th
during the off period has a negative r
given by:
iL(off)+(Vin* Vout)
L
The energy is then completely emp
capacitor and the switched terminal
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SMPSRM
When there is some residual energy permitted to
remain within the inductor core, the operation is called
continuous mode. This can be seen in Figure 3.
Energy for the entire on and off time periods must be
stored within the inductor. The stored energy is defined
by:
EL+ 0.5L @ ipk2 (eq. 7)
The boostmode inductor must store enough energy to
supply the output load for the entire switching period (ton+ toff). Also, boostmode converters are typically limited
to a 50 percent duty cycle. There m
when the inductor is permitted to
energy.
The boost converter is used fo
nonisolated) stepup applications a
than 100150 watts due to high pea
nonisolated converter, it is limited
less than 42.5 VDC. Replacing t
transformer results in a flyback conv
stepup or stepdown. The transfo
dielectric isolation from input to out
Vsat
Diode
ON
Vflbk(Vout)
Power
Switch
ON
Vin
Diode
ON
TIME
TIMEINDUCTORCURR
ENT
(AMPS)
SWITCHVOLTAGE
(VOLTS)
Figure 3. Waveforms for a ContinuousMode Boost Converter
Power
Switch
ON
Ipk
Common SwitchingPower Supply Topologies
A topology is the arrangement of the power devices
and their magnetic elements. Each topology has its own
merits within certain applications. There are five major
factors to consider when selecting a topology for a
particular application. These are:
1. Is inputtooutput dielectric isolation required forthe application? This is typically dictated by the
safety regulatory bodies in effect in the region.
2. Are multiple outputs required?
3. Does the prospective topology place a reasonable
voltage stress across the power semiconductors?
4. Does the prospective topology place a reasonable
5. How much of the input volta
the primary transformer win
Factor 1 is a safetyrelated issue. I
42.5 VDC are considered hazard
regulatory agencies throughout the
only transformerisolated topologies
this voltage. These are the offline ap
power supply is plugged into an AC
socket.
Multiple outputs require a
topology. The input and output
connected together if the input
42.5 VDC. Otherwise full dielectric
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Factors 3, 4 and 5 have a direct affect upon the
reliability of the system. Switching power supplies
deliver constant power to the output load. This power is
then reflected back to the input, so at low input voltages,the input current must be high to maintain the output
power. Conversely, the higher the input voltage, the
lower the input current. The design goal is to place as
much as possible of the input voltage across the
transformer or inductor so as to minimize the input
current.
Boostmode topologies have peak currents that are
about twice those found in forwardmode topologies.
This makes them unusable at output powers greater than100150 watts.
Cost is a major factor that enter
decision. There are large overlaps
boundaries between the topologies.
costeffective choice is to purposelyto operate in a region that usuall
another. This, though, may affect t
desired topology.
Figure 4 shows where the common
for a given level of DC input voltage
power. Figures 5 through 12 s
topologies. There are more topologi
as the Sepic and the Cuk, but they
used.
100010010
10
100
1000
OUTPUT POWER (W)
DCINPUTVOLTAGE(V)
42.5
Flyback
HalfBridge
FullBridge
Very High
Peak Currents
Buck
NonIsolated FullBridge
Figure 4. Where Various Topologies Are Used
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SMPSRM
Cout
Feedback
Power Switch
SW
Control
Control
Figure 5. The Buck (StepDown) Converter
Figure 6. The Boost (StepUp) Converter
VinCin
Vout
D
L
CinVin
VoutCout
+
+
+
+
+
D
L
Vin
Vin
IPK
0
IL
VFWD
0
VD
ILOAD IMIN
SW ON
D
O
D
ON
ID0
IL
VSAT
ISW
IPK
0
VSW
VFLBK
D
Feedback
SWControl
Figure 7. The BuckBoost (Inverting) Converter
CinVin
Cout Vout
+
+L
+
Vout
Vin
0
IL
0VL
IDISW
IPK
D+ Vin
0
SWON
VSAT
VSW
VFLBK
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Cout
Feedback
ControlSW
Figure 9. The OneTransistor Forward Converter (Half Forward Converter
VoutCinVin
N1 N2
TD+
+
+
LO
Cout
LO
Control
SW1
SW2
Feedback
Vout
Cin
Vin
T D1
D2
+
+
+
SW2
SW1VSW
2Vin
Vin
TIME0
TIME0
IPRI
IMIN
SWON
VSAT
VSW
2Vin
IPK
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SMPSRM
Feedback
VoutCout
TCin
C
C
LO
Control
Ds
+
+
+
N1
N2
SW2
SW1XFMR
Vin
0
SW1
SW2
VSAT
VSW2
TIME0
TIME
IPRI
IMIN
IPK
Vin
Vin2
Figure 11. The HalfBridge Converter
VoutCout
Vin
XFMR
Cin
LO
Control
SW1
SW2
Ds
XFMR
CN1
N2
TSW3
SW4
+
+
+
SW
1-4
SW
Vin
Vin2
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Interleaved Multiphase ConvertersOne method of increasing the output power of any
topology and reducing the stresses upon the
semiconductors, is a technique called interleaving. Any
topology can be interleaved. An interleaved multiphase
converter has two or more identical converters placed in
parallel which share key components. For an nphase
converter, each converter is driven at a phase difference
of 360/n degrees from the next. The output current from
all the phases sum together at the output, requiring only
Iout/n amperes from each phase.
The input and output capacitors a
phases. The input capacitor sees less
because the peak currents are less an
cycle of the phases is greater than iwith a single phase converter. The o
be made smaller because the fre
waveform is ntimes higher and its c
is greater. The semiconductors al
stress.
A block diagram of an interleav
converter is shown in Figure 13.
topology that is useful in providin
performance microprocessor.
Figure 13. Example of a TwoPhase Buck Converter with Voltage and Current Fe
+
VFDBK
Control
GND
CFA
CFB
GATEA1
GATEA2
GATEB2
GATEB1
SA1
SA2
SB2
+
LA
SB1 LB
VIN CIN
CS5308
Voltage Feedback
Current Feedback A
Current Feedback B
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SMPSRM
Selecting the Method of ControlThere are three major methods of controlling a
switching power supply. There are also variations of
these control methods that provide additional protectionfeatures. One should review these methods carefully and
then carefully review the controller IC data sheets to
select the one that is wanted.
Table 1 summarizes the features o
methods of control. Certain methods
certain topologies due to reasons of response.
Table 1. Common Control Methods Used in ICs
Control Method OC Protection Response Time Prefe
Average OC Slow Fo
o tage o ePulsebyPulse OC Slow Fo
Intrinsic Rapid B
urrent o eHysteretic Rapid Boost
Hysteric Voltage Average Slow Boost
Voltagemode control (see Figure 14) is typically used
for forwardmode topologies. In voltagemode control,
only the output voltage is monitored. A voltage error
signal is calculated by forming the difference between
Vout (actual) and Vout(desired). This error signal is thenfed into a comparator that compares it to the ramp voltage
generated by the internal oscillator section of the control
IC. The comparator thus converts the voltage error signal
into the PWM drive signal to the power switch. Since the
only control parameter is the output voltage, and there is
inherent delay through the power circuit, voltagemode
control tends to respond slowly to input variations.
Overcurrent protection for a voltagemode controlled
converter can either be based on the average outputcurrent or use a pulsebypulse method. In average
overcurrent protection, the DC output current is
monitored, and if a threshold is exceeded, the pulse width
of the power switch is reduced. In pulsebypulse
overcurrent protection, the peak current of each power
switch on cycle is monitored and the power switch is
instantly cutoff if its limits are ex
better protection to the power switc
Currentmode control (see Figure
with boostmode converters. Cur
monitors not only the output voltage
current. Here the voltage error sign
the peak current within the magne
each power switch ontime. Current
very rapid input and output respon
inherent overcurrent protection. It is
for forwardmode converters; their
have much lower slopes in their
which can create jitter within compa
Hysteretic control is a method of c
keep a monitored parameter betwee
are hysteretic current and voltage c
they are not commonly used.
The designer should be very caref
prospective control IC data sheet. Th
and any variations are usually not c
the first page of the data sheet.
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+
+
+
+
+
Cur.
Comp.
Volt
Comp.
OSC Charge
Clock Ramp
Discharge
Steering
AverageOvercurrentProtection
PulsewidthComparator
PulsebyPulseOvercurrentProtection
VCC
VSS
RCS
VerrorAmp.
Ct
Vref
VOC
Iout(lavOC)
or
ISW(PPOC)
Figure 14. VoltageMode Control
VFB
Current Amp.
+
+
VoltComp.
OSC
Discharge
VCC
VerrorAmp.
Ct
Vref
+
S
R
Q
S R S
VFB
CurrentComparator
Verror
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SMPSRM
The Choice of SemiconductorsPower Switches
The choice of which semiconductor technology to use
for the power switch function is influenced by manyfactors such as cost, peak voltage and current, frequency
of operation, and heatsinking. Each technology has its
own peculiarities that must be addressed during the
design phase.
There are three major power switch choices: the
bipolar junction transistor (BJT), the power MOSFET,
and the integrated gate bipolar transistor (IGBT). The
BJT was the first power switch to be used in this field and
still offers many cost advantages over the others. It is alsostill used for very low cost or in high power switching
converters. The maximum frequency of operation of
bipolar transistors is less than 80100 kHz because of
some of their switching characteristics. The IGBT is used
for high power switching converters, displacing many of
the BJT applications. They too, though, have a slower
switching characteristic which limits their frequency of
operation to below 30 kHz typically although some can
reach 100 kHz. IGBTs have smaller die areas than powerMOSFETs of the same ratings, which typically means a
lower cost. Power MOSFETs are used in the majority of
applications due to their ease of use and their higher
frequency capabilities. Each of the technologies will be
reviewed.
The Bipolar Power TransistorThe BJT is a current driven device. That means that the
base current is in proportion to the current drawn throughthe collector. So one must provide:
IBu IC hFE (eq. 8)
In power transistors, the average gain (hFE) exhibited at
the higher collector currents is between 5 and 20. This
could create a large base drive loss if the base drive circuit
is not properly designed.
One should generate a gate drive vo
to 0.7 volts as possible. This is to
created by dropping the base drive vo
base current to the level exhibited bA second consideration is thestora
the collector during its turnoff trans
is overdriven, or where the base c
needed to sustain the collector cu
exhibits a 0.32 ms delay in its
proportional to the base overdrive. A
time is not a major source of loss,
limit the maximum switching
bipolarbased switching power supmethods of reducing the storage tim
switching time. The first is to us
capacitor whose value, typically arou
in parallel with the base curren
(Figure 16a). The second is to use pr
(Figure 16b). Here, only the amou
current is provided by the drive cir
excess around the base into the colle
The last consideration with B
excessive second breakdown. Th
caused by the resistance of the b
permitting the furthest portions of the
later. This forces the current being
collector by an inductive load, to
opposite ends of the die, thus ca
localized heating on the die. Th
shortcircuit failure of the BJT
instantaneously if the amount of c
great, or it can happen later if the a
less. Current crowding is always
inductive load is attached to the col
the BJT faster, with the circuits in
greatly reduce the effects of second
reliability of the device.
VBB
Control IC
VBB
100 pF
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The Power MOSFETPower MOSFETs are the popular choices used as
power switches and synchronous rectifiers. They are, on
the surface, simpler to use than BJTs, but they have somehidden complexities.
A simplified model for a MOSFET can be seen in
Figure 17. The capacitances seen in the model are
specified within the MOSFET data sheets, but can be
nonlinear and vary with their applied voltages.
Coss
CDG
CGS
Figure 17. The MOSFET Model
From the gate terminal, there are t
designer encounters, the gate input c
the draingate reverse capacitance (C
capacitance is a fixed value causedformed between the gate metalizatio
Its value usually falls in the rang
depending upon the physical co
MOSFET. The Crss is the capacitanc
and the gate, and has values in the r
Although the Crss is smaller, it
pronounced effect upon the gate d
drain voltage to the gate, thus dump
into the gate input capacitance. Thwaveforms can be seen in Figure 1
only the Ciss being charged or
impedance of the external gate driv
shows the effect of the changing d
coupled into the gate through Crsobserve the flattening of the gate
this period, both during the turnon
MOSFET. Time period t3 is the a
voltage provided by the drive cirneeded by the MOSFET.
+
0
IG
0
VDS
VGS
0
TURN
ON
TURNOFFVDR
VthVpl
t3 t3
t2t1 t2 t1
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SMPSRM
The time needed to switch the MOSFET between on
and off states is dependent upon the impedance of the
gate drive circuit. It is very important that the drive circuit
be bypassed with a capacitor that will keep the drivevoltage constant over the drive period. A 0.1 mF capacitor
is more than sufficient.
Driving MOSFETs in SwitchingPower Supply Applications
There are three things that are ve
high frequency driving of MOSFEtotempole driver; the drive voltage
bypassed; and the drive devices mu
high levels of current in very short p
compliance). The optimal drive c
Figure 19.
Figure 19. Bipolar and FETBased Drive Circuits (a. Bipolar Drivers, b. MOSFET
LOADVG
Ron
a. Passive TurnON
LOVG
Roff
b. Passive TurnOFF
LOADVG
c. Bipolar Totempole
LOVG
d. MOS Totempole
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Sometimes it is necessary to provide a
dielectricallyisolated drive to a MOSFET. This is
provided by a drive transformer. Transformers driven
from a DC source must be capacitively coupled from thetotempole driver circuit. The secondary winding must
be capacitively coupled to the gate with a DC restoration
circuit. Both of the series capacitors
10 times the value of the Ciss of the M
capacitive voltage divider that is fo
capacitors does not cause an excesscircuit can be seen in Figure 20.
VG
1 k
C RGT
C
C > 10
Ciss
1:1
Figure 20. TransformerIsolated Gate Drive
The Insulated Gate BipolarTransistor (IGBT)
The IGBT is a hybrid device with a MOSFET as the
input device, which then drives a siliconcontrolled
rectifier (SCR) as a switched output device. The SCR is
constructed such that it does not exhibit the latching
characteristic of a typical SCR by making its feedback
gain less than 1. The die area of the typical IGBT is less
than onehalf that of an identically rated power MOSFET,which makes it less expensive for highpower converters.
The only drawback is the turnoff characteristic of the
IGBT. Being a bipolar minority carrier device, charges
must be removed from the PN junctions during a turnoff
condition. This causes a current tail at the end of the
turnoff transition of the current waveform. This can be a
significant loss because the voltage across the IGBT is
very high at that moment. This makes the IGBT useful
only for frequencies typically less than 20 kHz, or forexceptional IGBTs, 100 kHz.
To drive an IGBT one uses the MOSFET drive circuits
shown in Figures 18 and 19. Driving the IGBT gate faster
makes very little difference in the performance of an
IGBT, so some reduction in drive currents can be used.
RectifiersRectifiers represent about 60 per
nonsynchronous switching power su
has a very large effect on the effic
supply.
The significant rectifier parame
operation of switching power suppli
forward voltage drop (Vf), which
across the diode when a forward the reverse recovery time (trr), wh
requires a diode to clear the mino
its junction area and turn off whe
is applied
theforward recovery time (tfrr) w
take a diode to begin to conduct f
after a forward voltage is applied
There are four choices of rec
standard, fast and ultrafast recoverybarrier types.
A standard recovery diode is
5060 Hz rectification due to
characteristics. These include comm
the 1N4000 series diodes. Fastre
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SMPSRM
a high reverse leakage current. For a typical switching
power supply application, the best choice is usually a
Schottky rectifier for output voltages less than 12 V, and
an ultrafast recovery diode for all other output voltages.The major losses within output rectifiers are
conduction losses and switching losses. The conduction
loss is the forward voltage drop times the current flowing
through it during its conduction period. This can be
significant if its voltage drop and current are high. The
switching losses are determined by how fast a diode turns
off (trr) times the reverse voltage across the rectifier. This
can be significant for high output voltages and currents.
The characteristics of power r
applications in switching power sup
great detail in Reference (5).
The major losses within outconduction losses and switching los
loss is the forward voltage drop time
through it during its conduction p
significant if its voltage drop and cu
switching losses are determined by h
off (trr) times the reverse voltage acro
can be significant for high output vo
Table 2. Types of Rectifier Technologies
Rectifier Type Average Vf Reverse Recovery Time Typic
Standard Recovery 0.71.0 V 1,000 ns 506
Fast Recovery 1.01.2 V 150200 ns Out
UltraFast Recovery 0.91.4 V 2575 nsOutp
Schottky 0.30.8 V < 10 ns Out
Table 3. Estimating the Significant Parameters of the Power Semiconductors
Bipolar Pwr Sw MOSFET Pwr Swopo ogy
VCEO IC VDSS ID VR
Buck Vin Iout Vin Iout Vin
Boost Vout(2.0 Pout)Vin(min)
Vout(2.0 Pout)Vin(min)
Vout
Buck/Boost Vin* Vout(2.0 Pout)
Vin(min)Vin* Vout
(2.0 Pout)
Vin(min)Vin* V
Flyback 1.7 Vin(max)(2.0 Pout)
Vin(min)1.5 Vin(max)
(2.0 Pout)
Vin(min)5.0 Vo
1 Transistor
Forward2.0 Vin
(1.5 Pout)
Vin(min)2.0 Vin
(1.5 Pout)
Vin(min)3.0 Vo
PushPull 2.0 Vin(1.2 Pout)
Vin(min)2.0 Vin
(1.2 Pout)
Vin(min)2.0 Vo
HalfBridge Vin(2.0 Pout)
Vin(min)Vin
(2.0 Pout)
Vin(min)2.0 Vo
FullBridge Vi(1.2 Pout) Vi
(2.0 Pout) 2 0 V
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The Magnetic ComponentsThe magnetic elements within a switching power
supply are used either for steppingup or down a
switched AC voltage, or for energy storage. Inforwardmode topologies, the transformer is only used
for steppingup or down the AC voltage generated by the
power switches. The output filter (the output inductor
and capacitor) in forwardmode topologies is used for
energy storage. In boostmode topologies, the
transformer is used both for energy storage and to provide
a stepup or stepdown function.
Many design engineers consider the magnetic
elements of switching power supplies counterintuitiveor too complicated to design. Fortunately, help is at hand;
the suppliers of magnetic components have applications
engineers who are quite capable of performing the
transformer design and discussing the tradeoffs needed
for success. For those who are more experienced or more
adventuresome, please refer to Reference 2 in the
Bibliography for transformer design guidelines.
The general procedure in the design of any magnetic
component is as follows (Reference 2, p 42):1. Select an appropriate core material for the
application and the frequency of operation.
2. Select a core form factor that is appropriate for
the application and that satisfies applicable
regulatory requirements.
3. Determine the core crosssectional area
necessary to handle the required power
4. Determine whether an airgap is needed and
calculate the number of turns needed for eachwinding. Then determine whether the accuracy
of the output voltages meets the requirements
and whether the windings will fit into the
selected core size.
5. Wind the magnetic component using proper
winding techniques.
6. During the prototype stage, verify the
components operation with respect to the level
of voltage spikes, crossregulation, outputaccuracy and ripple, RFI, etc., and make
corrections were necessary.
The design of any magnetic component is a calculated
estimate. There are methods of stretching the design
limits for smaller size or lower losses, but these tend to
Coiltronics, Division of Cooper El
Technology
6000 Park of Commerce Blvd
Boca Raton, FL (USA) 33487website: http://www.coiltronics.c
Telephone: 5612417876
Cramer Coil, Inc.
401 Progress Dr.
Saukville, WI (USA) 53080
website: http://www.cramerco.co
email: techsales@cramercoil.com
Telephone: 2622682150
Pulse, Inc.
San Diego, CA
website: http://www.pulseeng.com
Telephone: 8586748100
TDK
1600 Feehanville Drive
Mount Prospect, IL 60056
website: http://www.component.tTelephone: 8478036100
Laying Out the Printed CThe printed circuit board (PCB)
critical portion of every switching p
in addition to the basic design and th
Improper layout can adversely af
component reliability, efficiency a
PCB layout will be different, b
appreciates the common factors pre
power supplies, the process will be
All PCB traces exhibit inductan
These can cause high voltage transit
is a high rate of change in current f
trace. For operational amplifiers s
power signals, it means that the
impossible to stabilize. For traces tha
the current flowing through them, it m
from one end of the trace to the oth
can be an antenna for RFI. In a
coupling between adjacent traces
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Within all switching power supplies, there are four
major current loops. Two of the loops conduct the
highlevel AC currents needed by the supply. These are
the power switch AC current loop and the output rectifierAC current loop. The currents are the typical trapezoidal
current pulses with very high peak currents and very
rapid di/dts. The other two current loops are the input
source and the output load current loops, which carry low
frequency current being supplied from the voltage source
and to the load respectively.
For the power switch AC current loop, current flows
from the input filter capacitor through the inductor or
transformer winding, through the power switch and backto the negative pin of the input capacitor. Similarly, the
output rectifier current loops current flows from the
inductor or secondary transformer winding, through the
rectifier to the output filter capaci
inductor or winding. The filter cap
components that can source and sin
AC current in the time needed by tsupply. The PCB traces should be m
short as possible, to minimize resi
effects. These traces should be the f
Turning to the input source and
loops, both of these loops must be c
their respective filter capacitors t
switching noise could bypass the fi
capacitor and escape into the enviro
called conducted interference. Thesin Figure 21 for the two major f
power supplies, nonisolated
transformerisolated (Figure 21b).
+
Cin Cout
Vin
L
Control
Input CurrentLoop
Join Join
Power SwitchCurrent Loop
Join
GNDAnalog
SW
Output LoadGround
Output RectifierGround
PowerSwitch Ground
Input SourceGround
Vout
VFB
Output RectifierCurrent Loop
Cin
Cout
VinControl
Input CurrentLoop
Power SwitchCurrent Loop
SW
OutpuCurre
OutpGrou
Output RectifierGround
V
Output RectifierCurrent Loop
R
(a) The NonIsolated DC/DC Converter
+
A B
C
B
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The grounds are extremely important to the proper
operation of the switching power supply, since they form
the reference connections for the entire supply; each
ground has its own unique set of signals which canadversely affect the operation of the supply if connected
improperly.
There are five distinct grounds within the typical
switching power supply. Four of them form the return
paths for the current loops described above. The
remaining ground is the lowlevel analog control ground
which is critical for the proper operation of the supply.
The grounds which are part of the major current loops
must be connected together exactly as shown inFigure 21. Here again, the connecting point between the
highlevel AC grounds and the input or output grounds
is at the negative terminal of the appropriate filter
capacitor (points A and B in Figures 21a and 21b). Noise
on the AC grounds can very easily escape into the
environment if the grounds are not directly connected to
the negative terminal of the filter capacitor(s). The
analog control ground must be connected to the point
where the control IC and associated circuitry mustmeasure key power parameters, such as AC or DC
current and the output voltage (point C in Figures 21a and
21b). Here any noise introduced by large AC signals
within the AC grounds will sum directly onto the
lowlevel control parameters and greatly affect the
operation of the supply. The purpose of connecting the
control ground to the lower side of the current sensing
resistor or the output voltage resistor divider is to form a
Kelvin contact where any common mode noise is notsensed by the control circuit. In short, follow the example
given by Figure 21 exactly as shown for best results.
The last important factor in the
layout surrounding the AC voltage n
drain of the power MOSFET (or col
the anode of the output rectifier(scapacitively couple into any trace o
the PCB that run underneath the A
mount designs, these nodes also need
to provide heatsinking for the powe
This is at odds with the desire to kee
possible to discourage capacitive
traces. One good compromise is to m
the AC node identical to the AC nod
with many vias (platedthrough h
increases the thermal mass of the
heatsinking and locates any surr
laterally where the coupling capacita
An example of this can be seen in F
Many times it is necessary to para
to reduce the amount of RMS r
capacitor experiences. Close attenti
this layout. If the paralleled capacit
capacitor closest to the source of th
operate hotter than the others, shor
life; the others will not see this leve
ensure that they will evenly share
ideally, any paralleled capacitors sh
radiallysymmetric manner around
typically a rectifier or power switch
The PCB layout, if not done prope
paper design. It is important to guidelines and monitor the layout
process.
Power Device
Via
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Losses and Stresses in SwitchingPower Supplies
Much of the designers time during a switching power
supply design is spent in identifying and minimizing thelosses within the supply. Most of the losses occur in the
power components within the switching power supply.
Some of these losses can also present stresses to the
power semiconductors which may affect the long term
reliability of the power supply, so knowing where they
arise and how to control them is important.
Whenever there is a simultaneous voltage drop across
a component with a current flowing through, there is a
loss. Some of these losses are controllable by modifying
the circuitry, and some are controlled
a different part. Identifying the major
be as easy as placing a finger on eac
in search of heat, or measuring the cassociated with each power com
oscilloscope, AC current probe and
Semiconductor losses fall int
conduction losses and switching los
loss is the product of the terminal
during the power devices on pe
conduction losses are the saturation
power transistor and the on loss o
shown in Figure 23 and Figure 24 re
TURN-ON
CURRENT
CURRENT
TAIL
TURN-OFF
CURRENT
SATURATION
CURRENT
PINCHING OFF INDUCTIVE
CHARACTERISTICS OF THETRANSFORMER
IPEAK
COLLECTOR
CURRENT
(AMPS)
FALL
TIME
STORAGE
TIME
DYNAMIC
SATURATION
RISE
TIME
SATURATIONVOLTAGE
VPEAK
COLLECTOR-TO-EMITTER
(VOLTS)
SATURATION
LOSSTURN-ON
LOSS
TURN-OFF LOSS
SWITCHING LOSSNSTANTANEOUSENE
RGY
LOSS(JOULES)
CURRENT
CROWDING
PERIODSECOND
BREAKDOWN
PERIOD
DRAIN-TO-SOURCEVOLTAGE
(VOLTS)
DRAINC
URRENT
(AM
PS)
INSTANTANEOUSE
NERGY
LOSS(JOULES)
RISE
TIME
ON VOLTAGE
VPEAK
TURN-ON
CURRENT
ON CURRENT
PINCHING OFF INDUCT
CHARACTERISTICS OF
TRANSFORMER
CLEARING
RECTIFIERS
ON LOSS
TURN-ON
LOSS
TUR
SWIT
CLEARING
RECTIFIERS
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The forward conduction loss of a rectifier is shown in
Figure 25. During turnoff, the rectifier exhibits a reverse
recovery loss where minority carriers trapped within the
PN junction must reverse their direction and exit thejunction after a reverse voltage is applied. This results in
what appears to be a current flowing in reverse through
the diode with a high reverse terminal voltage.
The switching loss is the instantaneous product of the
terminal voltage and current of a power device when it is
transitioning between operating states (ontooff and
offtoon). Here, voltages are transitional between
fullon and cutoff states while simultaneously the current
is transitional between fullon and cutoff states. This
creates a very large VI product whic
the conduction losses. Switching loss
frequency dependent loss within ev
power supply.The lossinduced heat generation
the power component. This can b
effective thermal design. For bipola
however, excessive switching losse
lethal stress to the transistor in t
breakdown and current crowding fai
taken in the careful analysis of each
BiasedSafe Operating Area (FB
BiasedSafe Operating Area (RBSO
Figure 25. Stresses and Losses within Rectifiers
REVERSE VOLTAGE
FORWARD VOLTAGE
DIODEVOLTAGE
(VOLTS)
DEGREE OF DIODE
RECOVERY
ABRUPTNESS REVERSE
RECOVERY
TIME (Trr)
FORWARD CONDUCTION CURRENT
FORWARD
RECOVERY
TIME (Tfr)
IPK
DIODECURRENT
(AMPS)
SWITCHING
LOSS
FORWARD CONDUCTION LOSS
INSTANTANEOUSENERGY
LOSS(JOULES)
Techniques to Improve Efficiency inSwitching Power Supplies
The reduction of losses is important to the efficient
rectification is a technique to reduce
by using a switch in place of the diod
rectifier switch is open when the pow
d l d h th it
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SMPSRM
the MOSFET should be placed in parallel with the
synchronous MOSFET. The MOSFET does contain a
parasitic body diode that could conduct current, but it is
lossy, slow to turn off, and can lower efficiency by 1% to2%. The lower turnon voltage of the Schottky prevents
the parasitic diode from ever conducting and exhibiting
its poor reverse recovery characteristic.
Using synchronous rectification, the conduction
voltage can be reduced from 400 mV to 100 mV or less.
An improvement of 15 percent can be expected for the
typical switching power supply.
The synchronous rectifier can be d
that is directly controlled from
passively, driven from other signacircuit. It is very important to provid
drive between the power switch(es)
rectifier(s) to prevent any shootthr
dead time is usually between 50 to 1
circuits can be seen in Figure 26.
LO
+
Vo
+
VoVin
SWDrive
GND
Direct
DC
RGC
1:1
C > 10 Ciss
SR
Primary
VG
1 k
(a) Actively Driven Synchronous Rectifiers
TransformerIsolated
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Snubbers and ClampsSnubbers and clamps are used for two very different
purposes. When misapplied, the reliability of the
semiconductors within the power supply is greatlyjeopardized.
A snubber is used to reduce the level of a voltage spike
and decrease the rate of change of a voltage waveform.
This then reduces the amount of overlap of the voltage
and current waveforms during a transition, thus reducing
the switching loss. This has its benefits in the Safe
Operating Area (SOA) of the semiconductors, and it
reduces emissions by lowering the spectral content of any
RFI.A clamp is used only for reducing the level of a voltage
spike. It has no affect on the dV/dt of the transition.
Therefore it is not very useful for
useful for preventing comp
semiconductors and capacitors from
breakdown.Bipolar power transistors suffer fro
which is an instantaneous failure mo
occurs during the turnoff voltage t
than 75 percent of its VCEO rating, i
current crowding stress. Here both t
the voltage and the peak voltage o
controlled. A snubber is needed to
within its RBSOA (Reverse Bias Sa
rating. Typical snubber and clamp cFigure 27. The effects that these hav
switching waveform are shown in F
Figure 27. Common Methods for Controlling Voltage Spikes and/or RFI
ZE
CLA
SOFT
CLAMP
SNUBBERSNUBBERSOFT
CLAMP
ZENER
CLAMP
SNUBBER
CLAMP
ORIGINAL
WAVEFORM
VOLTAGE(VOL
TS)
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The Lossless SnubberA lossless snubber is a snubber whose trapped energy
is recovered by the power circuit. The lossless snubber is
designed to absorb a fixed amount of energy from thetransition of a switched AC voltage node. This energy is
stored in a capacitor whose size dictates how much
energy the snubber can absorb. A typical implementation
of a lossless snubber can be seen in Figure 29.
The design for a lossless snubber varies from topology
to topology and for each desired transition. Some
adaptation may be necessary for each circuit. The
important factors in the design of a lossless snubber are:
1. The snubber must have initial conditions thatallow it to operate during the desired transition
and at the desired voltages. Lossless snubbers
should be emptied of their energy prior to the
desired transition. The voltage to which it is
reset dictates where the snubber will begin to
operate. So if the snubber is reset to the input
voltage, then it will act as a lossless clamp which
will remove any spikes above the input voltage.
2. When the lossless snubber is
energy should be returned to
capacitor or back into the ou
Study the supply carefully. Renergy to the input capacitor
to use the energy again on th
Returning the energy to grou
mode supply does not return
reuse, but acts as a shunt cur
the power switch. Sometime
transformer windings are us
3. The reset current waveform
limited with a series inductoadditional EMI from being g
2 to 3 turn spiral PCB induc
greatly lower the di/dt of the
lossless snubber.
+
VSW ID
Unsnubbed VSW
Snubbed VSW
Drain Current (ID)
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The Active ClampAn active clamp is a gated MOSFET circuit that allows
the controller IC to activate a clamp or a snubber circuit
at a particular moment in a switching power supplyscycle of operation. An active clamp for a flyback
converter is shown in Figure 30.
In Figure 30, the active clamp is reset (or emptied of its
stored energy) just prior to the turn
then disabled during the negative tra
Obviously, the implementation o
more expensive than other approacreserved for very compact power su
a critical issue.
GND
+
ISW VSW
VDR
+
ICL
Unclamped
Switch Voltage
(VSW)
Clamped Switch
Voltage (VSW)
Switch
Current (ISW)
Drive
Voltage (VDR)
Clamp
Current (ICL)
Discharge
Vin
Figure 30. An Active Clamp Used in a One Transistor Forward or a Flyback Con
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QuasiResonant Topologies
A quasiresonant topology is designed to reduce or
eliminate the frequencydependent switching losses
within the power switches and rectifiers. Switchinglosses account for about 40% of the total loss within a
PWM power supply and are proportional to the switching
frequency. Eliminating these losses allows the designer
to increase the operating frequency of the switching
power supply and so use smaller inductors and
capacitors, reducing size and weight. In addition, RFI
levels are reduced due to the controlled rate of change of
current or voltage.
The downside to quasiresonant designs is that they
are more complex than nonresonant topologies due to
parasitic RF effects that must be considered when
switching frequencies are in the 100
Schematically, quasiresonant to
modifications of the standard PW
resonant tank circuit is added to the pto make either the current or the vo
a half a sinusoid waveform. Since t
zero and ends at zero, the product
current at the starting and ending po
no switching loss.
There are two quasiresonant me
switching (ZCS) or zero voltage sw
is a fixed ontime, variable offtim
ZCS starts from an initial conditioswitch is off and no current is fl
resonant inductor. The ZCS qu
converter is shown in Figure 31.
VinCin
CRVSW
FEEDBACK
VoutCout
LO
ILR
CONTROL
Vin
POWER SWITCH
ON
SWITCH
TURN-OFF
VSW
ILR
LR
A ZCS QuasiResonant Buck Converter
IPK
D
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In this design, both the power switch and the catch
diode operate in a zero current switching mode. Power is
passed to the output during the resonant periods. So to
increase the power delivered to the load, the frequencywould increase, and vice versa for decreasing loads. In
typical designs the frequency can change 10:1 over the
ZCS supplys operating range.
The ZVS is a fixed offtime, variable ontime method
control. Here the initial condition occurs when the power
switch is on, and the familiar current ramp is flowing
through the filter inductor. The ZVS quasiresonant buck
converter is shown in Figure 32. Here, to control the
power delivered to the load, the amo
times are varied. For light loads, th
When the load is heavy, the frequenc
ZVS power supply, the frequency over the entire operating range of th
There are other variations on the
promote zero switching losses, su
PWM, full and halfbridge topologi
and resonant transition topologies.
treatment, see Chapter 4 in th
Cookbook (Bibliography reference
LOLR
CRVouCoutFEEDBACK
D
VI/P
CONTROLCin
Vin
ILOAD
IPK
0
ISW
ID
A ZVS QuasiResonant Buck Converter
Vin* Vout
LR) L
OV
inLR
0
POWER SWITCH
TURNS ON
Vin
VI/P
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Power Factor CorrectionPower Factor (PF) is defined as the ratio of real power
to apparent power. In a typical AC power supply
application where both the voltage and current aresinusoidal, the PF is given by the cosine of the phase
angle between the input current and the input voltage and
is a measure of how much of the current contributes to
real power in the load. A power factor of unity indicates
that 100% of the current is contributing to power in the
load while a power factor of zero indicates that none of
the current contributes to power in the load. Purely
resistive loads have a power factor of unity; the current
through them is directly proportional to the appliedvoltage.
The current in an ac line can be thought of as consisting
of two components: real and imaginary. The real part
results in power absorbed by the load while the imaginary
part is power being reflected back into the source, such
as is the case when current and voltage are of opposite
polarity and their product, power, is negative.
It is important to have a power factor as close as
possible to unity so that none of the delivered power is
reflected back to the source. Reflected power is
undesirable for three reasons:
1. The transmission lines or power cord will
generate heat according to the total current
being carried, the real part plus the reflected
part. This causes problems for the electric
utilities and has prompted various regulations
requiring all electrical equip
a low voltage distribution sy
current harmonics and maxim
2. The reflected power not wasresistance of the power cord
unnecessary heat in the sour
stepdown transformer), con
premature failure and consti
3. Since the ac mains are limite
by their circuit breakers, it is
the most power possible from
available. This can only hap
power factor is close to or eqThe typical AC input rectificatio
bridge followed by a large input filt
the time that the bridge diodes con
driving an electrolytic capacitor, a n
This circuit will only draw current
when the inputs voltage exceeds the
capacitor. This leads to very high cur
of the input AC voltage waveform a
Since the conduction periods of ththe peak value of the current can be 3
input current needed by the equipme
only senses average current, so it w
peak current becomes unsafe, as fo
areas. This can present a fire haz
distribution systems, these current
neutral line, not meant to carry this k
again presents a fire hazard.
Clarge
110/220
AC VOLTS IN
I
Power used
Power
not used
VOLTAGE
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A Power Factor Correction (PFC) circuit is a switching
power converter, essentially a boost converter with a very
wide input range, that precisely controls its input current
on an instantaneous basis to match the waveshape andphase of the input voltage. This represents a zero degrees
or 100 percent power factor and mimics a purely resistive
load. The amplitude of the input current waveform is
varied over longer time frames to maintain a constant
voltage at the converters output filter capacitor. This
mimics a resistor which slowly changes value to absorb
the correct amount of power to meet the demand of the
load. Short term energy excesses and deficits caused by
sudden changes in the load are supplemented by a bulkenergy storage capacitor, the boost converters output
filter device. The PFC input filter capacitor is reduced to
a few microfarads, thus placing a halfwave haversine
waveshape into the PFC converter.
The PFC boost converter can operate down to about
30 V before there is insufficient voltage to draw any more
significant power from its input. The converter then can
begin again when the input haversine reaches 30 V on the
next halfwave haversine. This greatly increases theconduction angle of the input rectifiers. The dropout
region of the PFC converter is then filtered (smoothed)
by the input EMI filter.
A PFC circuit not only ensures that no power is
reflected back to the source, it also eliminates the
high current pulses associated with conventional
rectifierfilter input circuits. Because heat lost in the
transmission line and adjacent circuits is proportional to
the square of the current in the line, short strong current
pulses generate more heat than a pur
the same power. The active powe
circuit is placed just following the AC
example can be seen in Figure 34.Depending upon how much power
there is a choice of three differen
modes. All of the schematics for the
the same, but the value of the PF
control method are different. For in
than 150 watts, a discontinuousmo
typically used, in which the PFC
emptied prior to the next power swit
For powers between 150 and 250conduction mode is recommended.
control where the control IC senses
core is emptied of its energy and th
conduction cycle is immediately be
any dead time exhibited in the disc
control. For an input power greater
continuousmode of control is reco
peak currents can be lowered by
inductor, but a troublesome characteristic of the output rectif
which can add an additional 2040 pe
PFC circuit.
Many countries cooperate in the c
power factor requirements. The
document is IEC6100032, whic
performance of generalized electro
are more detailed specifications for
made for special markets.
Input Voltage
Switch Current
Conduction Angle
Figure 34. Power Factor Correction Circuit
Control
Csmall
I
Vsense
SMPSRM
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SMPSRM
Bibliography
1. BenYaakov Sam, Gregory Ivensky, Passive Lossless Snubbers for High Frequency PWM
Seminar 12, APEC 99.
2. Brown, Marty, Power Supply Cookbook, ButterworthHeinemann, 1994, 2001.
3. Brown, Marty, Laying Out PC Boards for Embedded Switching Supplies,Electronic Desi
4. Martin, Robert F., Harmonic Currents, Compliance Engineering 1999 Annual Resources
Communications, LLC, pp. 103107.
5. ON Semiconductor,Rectifier Applications Handbook, HB214/D, Rev. 2, Nov. 2001.
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SWITCHMODE Power Supply ExamplesThis section provides both initial and detailed information to simplify the selection and de
SWITCHMODE power supplies. The ICs for Switching Power Supplies figure identifies controloutput protection and switching regulator ICs for various topologies.
ICs for Switching Power Supplies . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Integrated circuits identified for various sections of a switching power supply.
Suggested Components for Specific Applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
A list of suggested control ICs, power transistors and rectifiers for SWITCHMODE power suppli
CRT Display System . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
AC/DC Power Supply for CRT Displays . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
AC/DC Power Supply for Storage, Imaging & Entertainment . . . . . . . . . . . . . . . . . . . . . . . . . . . .
DCDC Conversion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Typical PC ForwardMode SMPS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Real SMPS Applications
80 W Power Factor Correction Controller . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Compact Power Factor Correction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Monitor PulsedMode SMPS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
70 W Wide Mains TV SMPS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
100 W Wide Mains TV SMPS with 1.3 W Standby . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
LowCost Offline IGBT Battery Charger . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
110 W Output Flyback SMPS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Efficient Safety Circuit for Electronic Ballast . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
ACDC Battery Charger Constant Current with Voltage Limit . . . . . . . . . . . . . . . . . . . . . . . .
Some of these circuits may have a more complete application note, spice model information or evenavailable. Consult ON Semiconductors website (http://onsemi.com) or local sales office for more
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.com
36
Figure 30. Integrated Circuits for Switching P
MC33260
1N62xxA
MBR1100
TL4
MMBZ52xx
MMSZ52xx
MMSZ46xx
MC33362
HV SWITCHING
OUTPUT FILTERS
SNUBBER/
POWER FACTOR
POWER FACTOR
POWER
TRANS
VOLTCONTROLSTARTUP
CLAMP
CORRECTION
SNUBBER/CLAMP
OUTPUTFILTERSCORRECTION
FEEDB
REGULATORS
CONTROLSTARTUP
VOFEE
SWITCH
REF
PWM
OSC
FORMERS
POWER MOSDRIVERS
MC33151
MC33152
MC33153
POWER MOSDRIVERS
MC33363
MC33365NCP100x
NCP105x
CS51021CS51022CS51023CS51024CS5106CS51220CS51221
CS51227CS5124
MC33023MC33025MC33065MC33067MC33364
MC44603A
MC44604MC44605MC44608NCP1200NCP1205
TL
CS
NC
MBR3100
MBR360
MBRD360
MBRS1100
MBRS2
MBRS3
MURHF8
MURS
1N63xxAMUR160MUR260
MURS160MURS260P6KExxxA
P6SMB1xxA
MC33262
MC33368
MC34262
NCP1651
NCP1650
CS3843
UC384x
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On Screen Display
RGB
Ove
Geometry Correction
IRF630 / 640 / 730 /740 / 830 / 840
Timebase Processor
R
MEMORY
1280
V_Sync
DOWN
UP
USB HUB
USB & Auxiliary Standby
Line
PFC Devices
S.M.P.S
UC384xSync
V_Sync
H_Sync
MonitorMCU
SYNC PROCESSOR
RWM
10101100101
x1024
HC05CPU
CORE
AC/DC
Power Supply
RGB
A.C.
MC33363A/B
Controller
NCP1650NCP1651
V_Sync
BG
R
MC44603/5Signal
MUR420MUR440MUR460
Generator
H_Sync
H_Sync
Figure 31. 15 Monitor Power Sup
Figure3
7.
.15
Monitor
Power
Supplie
s
MC34262
MC44608
I2C BUS
PWM
or I2C
600V 8ANCh
MOSFET
MC33368
MC33260
NCP100xNCP105xNCP1200
NCP1200NCP1205
SMPSRM
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SMPSRM
Rectifier
ACLine
Bulk+
StorageCapacitor
StartupSwitch
PWMControl
IC
Prog.Prec.Ref
+Ultrafast
MOSFET
Rectifier
PWM Switcher
Figure 38. AC/DC Power Supply for CRT Displays
Table 1.
Part # Description Key Parameters
MC33262 PFC Control IC Critical Conduction PFC Controller
MC33368 PFC Control IC Critical Conduction PFC Controller + Internal Startup
MC33260 PFC Control IC Low System Cost, PFC with Synchronization
Capability, Follower Boost Mode, or Normal Mode
MC33365 PWM Control IC Fixed Frequency Controller + 700 V Startup, 1 A
Power Switch
MC33364 PWM Control IC Variable Frequency Controller + 700 V Startup Switch
MC44603A/604 PWM Control IC GreenLine, Sync. Facility with Low Standby Mode
MC44605 PWM Control IC GreenLine, Sync. Facility, Currentmode
MC44608 PWM Control IC GreenLine, Fixed Frequency (40 kHz, 75 kHz and 100
kHz options), Controller + Internal Startup, 8pin
MSR860 Ultrasoft Rectifier 600 V, 8 A, trr = 55 ns, Ir max = 1 uAMUR440 Ultrafast Rectifier 400 V, 4 A, trr = 50 ns, Ir max = 10 uA
MRA4006T3 Fast Recovery Rectifier 800 V, 1 A, Vf = 1.1 V @ 1.0 A
MR856 Fast Recovery Rectifier 600 V, 3 A, Vf = 1.25 V @ 3.0 A
NCP1200 PWM CurrentMode Controller 110 mA Source/Sink, O/P Protection, 40/60/110 kHz
NCP1205 SingleEnded PWM Controller Quasiresonant Operation, 250 mA Source/Sink,
836 V Operation
UC3842/3/4/5 High Performance CurrentMode
Controllers
500 kHz Freq., Totem Pole O/P, CyclebyCycle
Current Limiting, UV Lockout
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Rectifier
ACLine
Bulk+
StorageCapacitor
StartupSwitch
PWMControl
IC
Prog.Prec.Ref
+Ultrafast
MOSFET
Rectifier
PWM Switcher
Figure 39. AC/DC Power Supply for Storage,
Imaging & Entertainment
Table 2.
Part # Description Key Parameters
MC33363A/B/65 PWM Control IC Controller + 700 V Startup & Power Switch, < 15 W
MC33364 PWM Control IC Critical Conduction Mode, SMPS Controller
TL431B Program Precision Reference 0.4% Tolerance, Prog. Output up to 36 V, Temperatu
Compensated
MSRD620CT Ultrasoft Rectifier 200 V, 6 A, trr = 55 ns, Ir max = 1 uA
MR856 Fast Recovery Rectifier 600 V, 3 A, Vf = 1.25 V @ 3.0 A
NCP1200 PWM CurrentMode Controller 110 mA Source/Sink, O/P Protection, 40/60/110 kHz
NCP1205 SingleEnded PWM Controller Quasiresonant Operat ion, 250 mA Source/Sink,
836 V Operation
UC3842/3/4/5 High Performance CurrentMode
Controllers
500 kHz Freq., Totem Pole O/P, CyclebyCycle
Current Limiting, UV Lockout
SMPSRM
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+
Vin Control IC
+
Vout LoadCo
Lo
+
Vin
VoltageRegulation
Buck Regulator Synchronous Buck Re
Control IC
Lo
Figure 40. DC DC Conversion
Table 3.
Part # Description Key Parameters
MC33263 Low Noise, Low Dropout
Regulator IC
150 mA; 8 Outputs 2.8 V 5 V; SOT 23L 6 Lead
Package
MC33269 Medium Dropout Regulator IC 0.8 A; 3.3; 5, 12 V out; 1 V diff; 1% Tolerance
MC33275/375 Low Dropout Regulator 300 mA; 2.5, 3, 3.3, 5 V out
LP2950/51 Low Dropout, Fixed Voltage IC 0.1 A; 3, 3.3, 5 V out; 0.38 V diff; 0.5% Tolerance
MC78PC CMOS LDO Linear Voltage
Regulator
Iout= 150 mA, Available in 2.8 V, 3 V, 3.3 V, 5 V; SOT
23 5 Leads
MC33470 Synchronous Buck Regulator IC Digital Controlled; Vcc= 7 V; Fast Response
NTMSD2P102LR2 PCh FET w/Schottky in SO8 20 V, 2 A, 160 mW FET/1 A, Vf = 0.46 V Schottky
NTMSD3P102R2 PCh FET w/Schottky in SO8 20 V, 3 A, 160 mW FET/1 A, Vf = 0.46 V Schottky
MMDFS6N303R2 NCh FET w/Schottky in SO8 30 V, 6 A, 35 mW FET/3 A, Vf = 0.42 V Schottky
NTMSD3P303R2 PCh FET w/Schottky in SO8 30 V, 3 A, 100 mW FET/3 A, Vf = 0.42 V Schottky
MBRM140T3 1A Schottky in POWERMITE
Package
40 V, 1 A, Vf = 0.43 @ 1 A; Ir = 0.4 mA @ 40 V
MBRA130LT3 1A Schottky in SMA Package 40 V, 1 A, Vf = 0.395 @ 1 A; Ir = 1 mA @ 40 V
MBRS2040LT3 2A Schottky in SMB Package 40 V, 2 A, Vf = 0.43 @ 2 A; Ir = 0.8 mA @ 40 V
MMSF3300 Single NCh MOSFET in SO8 30 V, 11.5 A(1),12.5 mW @ 10 V
NTD4302 Single NCh MOSFET in DPAK 30 V, 18.3 A(1),10 mW @ 10 V
NTTS2P03R2 Single PCh MOSFET in
Micro8 Package
30 V, 2.7 A, 90 mW @ 10 V
MGSF3454X/V Single NCh MOSFET in
TSOP6
30 V, 4.2 A, 65 mW @ 10 V
NTGS3441T1 Single PCh MOSFET in
TSOP6
20 V, 3.3 A, 100 mW @ 4.5 V
NCP1500 Dual Mode PWM Linear Buck
Converter
Prog. O/P Voltage 1.0, 1.3, 1.5, 1.8 V
NCP1570 Low Voltage Synchronous Buck UV Lockout, 200 kHz Osc. Freq., 200 ns Response
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41
1N5404RL
V (V)Part No. RRM I (A)o
400 1000 3
Package
Axial
MBR160
Part No. I (A)o
60 1
Package
Axial
X
MUR180E, MUR1100E
V (V)
MUR480E, MUR4100EMR756RL, MR760RL1N4937 600
Part No. RRM I (A)o
600 1000461
1
Package
AxialAxialAxial
X600 1000X600 1000X
Axial
Mains230 Vac
+
+
+
+
PWM
MA
IC
V (V)RRM
+
Figure 35. Typical 200 W ATX Forward
VoltageStandby 5 V 0.1 A
U384X SeriesMC34060
TL494TL594MC34023
MC44608
Part No. Package
DIP14/SO14
DIP16/SO16DIP16/SO16DIP16/SO16
DIP8
DIP8/SO8/SO14
MC44603MC44603A
DIP16/SO16DIP16/SO16
+1N5406RL 400 1000 3 AxialX1N5408RL 400 1000 3 AxialX
SMPSRM
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Application: 80 W Power Factor Controller
0.01
C2
MULTIPLIER7.5 k
R33
1
+
+
100
C4
13 V/
8.0 V
500
N
MO
Q
2.2 M
R5
++
C5
D4
D3
D2
D1 6.7 V
ZERO CURRENT
DETECTOR
UVLO
CURRENTSENSE
COMPARATOR
RS
LATCH
1.2 V
1.6 V/
1.4 V
36 V
2.5 V
REFERENCE
16 V10
DRIVE
OUTPUT10
TIMER
DELAY
92 to
138 Vac
1
22 k
R4
100 k
R6 1N4D
0.1
R7
4
7
8
5
6 2
R
0.68
C1
1.5 V
ERROR AMP
OVERVOLTAGE
COMPARATOR
+1.08 Vref
+Vref
QUICKSTART
10 mA
10 pF
20 k
FILTER
RFI
Figure 42. 80 W Power Factor Controller
MC33262
Features:
Reduced part count, lowcost solution.
ON Semiconductor Advantages:
Complete semiconductor solution based around highly integrated MC33262.
Devices:
Part Number Description
MC33262 Power Factor Controller
MUR130 Axial Lead Ultrafast Recovery Rectifier (300 V)
Transformer Coilcraft N2881A
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Application: Compact Power Factor Correction
MAINS
FILTER
1
2
3
4
8
7
6
5
MC33260
100 nFAC LINE
FUSE 0.33 F
1N5404
Vcc
100 nF
12 kW
120 pF
0.5 W/3 W
10 F/
16 V
+
10 W
45 kW 1 MW
1 MW
L1
MUR460
500 V/8 A
NCh
MOSFET
Figure 43. Compact Power Factor Correction
Features :
Lowcost system solution for boost mode follower.Meets IEC100032 standard.
Critical conduction, voltage mode.
Follower boost mode for system cost reduction smaller inductor and MOSFET can be used.
Inrush current detection.
Protection against overcurrent, overvoltage and undervoltage.
ON Semiconductor advantages:
Very low component count.
No Auxiliary winding required.
High reliability.
Complete semiconductor solution.
Significant system cost reduction.
Devices:
SMPSRM
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Application: Monitor PulsedMode SMPS
90 Vac to
270 Vac
RFI
FILTER
D1 D4
1N5404 150 F
400 V
47 F
25 V
1N4934
100 nF
MR8561 H
SYNC
2.2 nF
10 pF9
10
11
12
13
14
15
16
8
7
6
5
4
3
2
1
0.1 W
470 pF
Lp
MBR3
4700
3.9 kW/6 W
TL431
MOC8107
1 nF/1 kV
4.7MW
MC44605P
MR85
220
1N4742A
12 V
MR85
1000
MR85
1000
2.7 kW
MR85
47
4.7 k
1N4934
100 W
47 kW
Note 1
270 W
10 W
560 kW4.7 F
10 V
1.8 MW
10 kW
1 kW
56 kW
1N4934
150 kW
4.7 F
10 V
Vin
+
56 kW
470
kW1N4148
1.2 kW
2.2 nF
2.2 kW
+
22
nF
4.7 F
3.3 kW
22 kW
2W
SMT31
8.2 kW+
1 W
+
1 nF/500 V
1 nF/500 V
Vin
470 pF
470 W
Note 1: 500 V/8 A NChannel MOSFET
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Features:
Off power consumption: 40 mA drawn from the 8 V output in Burst mode.
Vac (110 V) about 1 watt
Vac (240 V) about 3 wattsEfficiency (pout = 85 watts)
Around 77% @ Vac (110 V)
Around 80% @ Vac (240 V)
Maximum Power limitation.
Overtemperature detection.
Winding short circuit detection.
ON Semiconductor Advantages:
Designed around high performance current mode controller.
Builtin latched disabling mode.
Complete semiconductor solution.
Devices:
Part Number Description
MC44605P High Safety Latched Mode GreenLinet Controller
For (Multi) Synchronized Applications
TL431 Programmable Precision Reference
MR856 Fast Recovery Rectifier (600 V)
MR852 Fast Recovery Rectifier (200 V)
MBR360 Axial Lead Schottky Rectifier (60 V)
BC237B NPN Bipolar Transistor
1N5404 GeneralPurpose Rectifier (400 V)
1N4742A Zener Regulator (12 V, 1 W)
Transformer G635100 (SMT31M) from Thomson OregaPrimary inductance = 207 mH
Area = 190 nH/turns2
Primary turns = 33
Turns (90 V) = 31
SMPSRM
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Application: 70 W Wide Mains TV SMPS
95 Vac to265 Vac
RFI
FILTER
C4C51 nF/1 kV
D1D4
1N4007C1
220 mF
R7
68 kW/1 W
C16
100 F
D13
1N4148C26
4.7 nF
D7
1N4937L1
1 H
C8 560 pF
C10 1 F
R15
1 MW
C7
10 nF
R5
2.2 kW
R14
47 kW
R13
10 kW
R9 150W
15 kW
1 kW
R19
27kW
C121 nF
9
10
11
12
13
14
15
16
8
7
6
5
4
3
2
1
R8
1 kW
Q1
600 V/4 A
NCh
MOSFET
C14
220 pF
D12
MR856
D5
MR854
D8
MR854
C15 220 pF
C20
47 F
R16
68 kW/2 W
C191 nF/1 kV
R21
4.7MW
MC44603AP
R20 47W
R33
0.31 W
R4
3.9 kW
5.6 kW
LF1
C9
100 nF
D15
1N4148
C11
100 pF
3.8 MW
F1
FUSE 1.6 A
R18
OREGA TRANS
G6191THOMSON TV CO
R22
R3
22 kW
C30
100 nF
250 Vac
180 kW
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Features:
70 W output power from 95 to 265 Vac.
Efficiency
@ 230 Vac = 86%@ 110 Vac = 84%
Load regulation (115 Vac) =" 0.8 V.Cross regulation (115 Vac) =" 0.2 V.Frequency 20 kHz fully stable.
ON Semiconductor Advantages:
DIP16 or SO16 packaging options for controller.
Meets IEC emi radiation standards.
A narrow supply voltage design (80 W) is also available.Devices:
Part Number Description
MC44603AP Enhanced Mixed Frequency Mode
GreenLinet PWM Controller
MR856 Fast Recovery Rectifier (600 V)
MR854 Fast Recovery Rectifier (400 V)
1N4007 General Purpose Rectifier (1000 V)
1N4937 General Purpose Rectifier (600 V)
Transformer Thomson Orega SMT18
SMPSRM
A li i Wid M i 100 W TV SMPS i h 1 3 W TV S d
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Application: Wide Mains 100 W TV SMPS with 1.3 W TV Stand
RFI
FILTER
C4
1 nF
D1D4
1N5404C5
220 mF
400 V
C6
47 nF
630 VD6
MR856
D7
1N4148
Isense
R2
10 W
Vcc
1
2
3
4
8
7
6
5
D14
MR856
D18 MR856
D9 MR852
D10
MR852
C11
220 pF/500 V
C15
1000 F/16 V
C12
47 F/250 V
R1
22 kW
5W
C19
2N2FY
R16 4.7MW/4 kV
M
C44608P75
R4 3.9 kW
R30.27 W
47283900 R F6
F1C31
100 nF
C3
1 nF
R17
2.2 kW
5 W
C8
100 nF
R5 100 kW
1
2
8
7
6
11
10
9
D5
1N4007
OPT1
+ C7
22 mF
16 V
14
12
C13
100 nF
+
R7 47 k C17 120 pF
C14
1000 F/35 V
+
D12
1N4934
DZ1
MCR226
+
C9
470 pF
630 V
R21 47 W
+
C16
100 pF
R19
18 kW
R9
100 kW
R10
R12
1 kW
ON
O
600 V/6 A
NCH
MOSFET
F t
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Features:
Off power consumption: 300mW drawn from the 8V output in pulsed mode.
Pin = 1.3W independent of the mains.
Efficiency: 83%
Maximum power limitation.
Overtemperature detection.
Demagnetization detection.
Protection against open loop.
ON Semiconductor Advantages:
Very low component count controller.
Fail safe open feedback loop.
Programmable pulsedmode power transfer for efficient system standby mode.
Standby losses independent of the mains value.Complete semiconductor solution.
Devices:
Part Number Description
MC44608P75 GreenLinet Very High Voltage PWM Controller
TL431 Programmable Precision Reference
MR856 Fast Recovery Rectifier (600 V)
MR852 Fast Recovery Rectifier (200 V)
1N5404 General Purpose Rectifier (400 V)
1N4740A Zener Regulator (10 V, 1 W)
Transformer SMT19 4034629 (9 slots coil former)
Primary inductance: 181 mH
Nprimary: 40 turns
N 112 V: 40 turns
N 16 V: 6 turns
N 8 V: 3 turns
SMPSRM
Application: Low Cost Offline IGBT Battery Charger
7/22/2019 Con Mutada Son
51/65
Application: LowCost Offline IGBT Battery Charger
Figure 47. LowCost Offline IGBT Battery Charger
1N4148
D1R1
150
R3
220 k
C7
10 mF
C3
10 mF/
350 V
D2
12 V
R1
120 k
R5
1.2 k
C9
1 nF
Q1
MBT3946DW
R2
3.9C5
1 nF
R13
100 k
C10
1 nF
1N4937
D4
R9
470
MC14093
R5
1 k
8 7 6 5
1 2 3 4
R9